1. Technical Field
The present invention relates to a radar apparatus which detects a target by receiving a signal of reflection waves reflected by the target with an antenna.
2. Description of the Related Art
Radar apparatuses are devices for measuring a distance between a target and a measuring place, a direction of the target from the measuring place, and other items by radiating radio waves to the space from the measuring place and receiving a signal of reflection waves reflected by the target. In particular, in recent years, radar apparatuses which can detect targets including not only automobiles but also pedestrians etc. by a high-resolution measurement using short-wavelength radio waves such as microwaves or millimeter waves have been being developed.
There may occur a case that a radar apparatus receives a signal that is a mixture of reflection waves coming from a nearby target and reflection waves coming from a distant target. In particular, where range sidelobes occur due to the autocorrelation characteristic of a signal of reflection waves coming from a nearby target, a reception signal of a radar apparatus includes these range sidelobes and a main lobe of reflection waves coming from a distant target in mixture, as a result of which the accuracy of detection of the distant target is lowered in the radar apparatus.
When an automobile and a pedestrian are located at the same distance from a measuring place, a radar apparatus may receive a signal that is a mixture of signals of reflection waves coming from the automobile and the pedestrian which have different radar cross sections (RCSs). It is said that in general the radar cross section of a pedestrian (human) is smaller than that of an automobile. Therefore, radar apparatuses are required to properly receive not only reflection waves coming from an automobile but also reflection waves coming from a pedestrian even if they are located at the same distance from a measuring place.
Therefore, radar apparatuses which are required to be able to perform a high-resolution measurement on plural targets as mentioned above are also required to transmit pulse waves or pulse modulation waves having such a characteristic that range sidelobe levels are made lower (hereinafter referred to as a low range sidelobe characteristic). Such radar apparatuses are further required to have a wide reception dynamic range for a reception signal.
In connection with the above-described low range sidelobe characteristic, pulse compression radars are known which transmits pulse waves or pulse modulation waves which use a complementary code and have a low range sidelobe characteristic. The pulse compression is a method of transmitting a signal having a large pulse width obtained by pulse-modulating or phase-modulating a pulse signal and demodulating a received signal into a signal having a narrow pulse width in post-reception signal processing. The pulse compression can make it possible to increase the target detection distance and increase the detection distance estimation accuracy.
The complementary code is a code which comprises plural (e.g., two) complementary code sequences (an, bn) (parameter n=1, 2, . . . , L). The complementary code has a property that range sidelobes are made zero when autocorrelation calculation results, equalized in a shift time τ, of the one complementary code sequence an and the other complementary code sequence bn are added together. The parameter L represents a code sequence length or merely a code length.
A complementary code generating method will be disclosed with reference to FIG. 13. FIG. 13 is an explanatory diagram showing a procedure for generating general complementary code sequences. As shown in FIG. 13, subcode sequences (c, d) having elements “1” and “−1” and a pulse code length L=2p−1 are generated according to the statements on lines 4 and 5 and complementary code sequences (a, b) having a pulse code length L=2p are further generated according to the statements on lines 6 and 7. The one complementary code sequence a is a connection of the subcode sequences c and d. The other complementary code sequence b is a connection of the subcode sequences c and −d.
The symbol (a, b) represents the complementary code sequences and (c, d) represents the subcode sequences constituting each complementary code sequence. The parameter p determines the code length L of the generated complementary code sequences (a, b).
Characteristics of the above complementary code will be described with reference to FIG. 14. FIG. 14 is explanatory diagrams showing characteristics of a conventional complementary code. In FIG. 14, (a) is an explanatory diagram showing an autocorrelation calculation result of the one complementary code sequence an. In FIG. 14, (b) is an explanatory diagram showing an autocorrelation calculation result of the other complementary code sequence bn. In FIG. 14, (c) is an explanatory diagram showing addition values of the autocorrelation calculation results of the two respective complementary code sequences (an, bn). The code length L of the complementary code shown in FIG. 14 is equal to 128.
Between the two complementary code sequences (an, bn), an autocorrelation calculation result of the one complementary code sequence an is calculated according to Equation (1). An autocorrelation calculation result of the other complementary code sequence bn is calculated according to Equation (2). The parameter R represents an autocorrelation calculation result. It is assumed that each of the complementary code sequences an and bn is zero when n>L or n<1 (i.e., an=0 and bn=0 when n>L or n<1). The asterisk * is a complex conjugate operator.
                    [                  Formula          ⁢                                          ⁢          1                ]                                                                                  R            aa                    ⁡                      (            τ            )                          =                              ∑                          n              =              1                        L                    ⁢                                    a              n                        ⁢                          a                              n                +                τ                            *                                                          (        1        )                                [                  Formula          ⁢                                          ⁢          2                ]                                                                                  R            bb                    ⁡                      (            τ            )                          =                              ∑                          n              =              1                        L                    ⁢                                    b              n                        ⁢                          b                              n                +                τ                            *                                                          (        2        )            
As shown in (a) in FIG. 14, the autocorrelation calculation result Raa(τ) of the one complementary code sequence an calculated according to Equation (1) has a peak when the shift time τ is equal to 0 and has range sidelobes for the shift time τ being not equal to 0. Likewise, as shown in (b) in FIG. 14, the autocorrelation calculation result Rbb(τ) of the other complementary code sequence bn calculated according to Equation (2) has a peak when the shift time τ is equal to 0 and has range sidelobes for the delay time τ being not equal to 0.
As shown in (c) in FIG. 14, the addition values of the autocorrelation calculation results Raa(τ) and Rbb(τ) have a peak (in the following description, a peak occurring when the shift time τ is equal to 0 will be referred to as a main lobe) when the shift time (delay time) τ is equal to 0 and have no range sidelobes (i.e., have values 0) for the shift time τ being not equal to 0. This is expressed as Formulae (3). In FIGS. 14(a)-14(c), the horizontal axis represents the shift time τ which is used in the autocorrelation calculation and the vertical axis represents the calculated autocorrelation calculation result.[Formulae 3]Raa(τ)+Rbb(τ)≠0, when τ=0Raa(τ)+Rbb(τ)=0, when τ≠0  (3)
Next, a reception dynamic range of a pulse compression radar as an example of the above kind of radar apparatus when plural targets are detected by the pulse compression radar will be described with reference to FIG. 15. FIG. 15 is a conceptual diagram illustrating a reception dynamic range of a radar receiving unit when plural targets TR1, TR2, and TR3 are detected by a conventional pulse compression radar apparatus 1z. As shown in FIG. 15, the pulse compression radar apparatus 1z transmits a transmission signal having a pulse width Tp and a pulse compression code length L in such a manner as to continue transmission of a pulse sequence during a pulse sequence transmission interval T. Equation (4) holds between the pulse sequence transmission interval T, the pulse width Tp, and the pulse compression code length L:[Formula 4]T=Tp×L  (4)
As shown in FIG. 15, assume that a target TR2 exists at a position having a distance R from the pulse compression radar apparatus 1z and targets TR1 and TR3 exist within a distance range [R−(cT/2), R+(cT/2)]. In this case, a (reflection) signal RS2 which is reflection waves coming from the target TR2 overlaps with (reflection) signals RS1 and RS3 which are reflection waves coming from the other targets TR1 and TR3 existing in the distance range [R−(cT/2), R+(cT/2)]. The parameter c is the speed of light (m/s).
Therefore, to suppress degradation of the distance measurement accuracy of the pulse compression radar apparatus 1z, a reception dynamic range is necessary which enables proper reception of each of the reception wave signals RS1-RS3 which come from the respective targets TR1-TR3 and include overlap portions. If the pulse compression radar apparatus 1z does not have a proper reception dynamic range, the peak level lowers and the range sidelobe level increases when pulse compression is done, resulting in reduction in the measurement accuracy of distances of the targets TR1-TR3.
In measurements by conventional radar apparatuses, the distance propagation loss is proportional to the forth power of the distance. Therefore, as the pulse compression code length L becomes greater, the processing gain of the pulse compression increases and the measurable distance range increases. However, as the pulse compression code length L becomes greater, the reception dynamic range that is necessary for reception increases. The radar equation produces a conclusion that a dynamic range that is necessary for reception of signals coming from plural nearby targets (within about 30 m) is wider than a dynamic range that is necessary for reception of signals coming from plural distant targets (more distant than about 30 m).
The distributed compression type pulse echo system transceiver disclosed in Patent document 1 is known as a device which relates to the above-described reception dynamic range, more specifically, a device which suppresses the reception dynamic range while increasing the measurable distance range.
This distributed compression type pulse echo system transceiver transmits, time-divisionally, high-frequency signals modulated according to pulse compression codes that are code sequences having different code lengths, respectively, in respective modes (B mode and Doppler mode). More specifically, in the B mode, this transceiver transmits a high-frequency signal for a short-distance range that is modulated according to a code (code sequence) having a short pulse code length. In the Doppler mode, this transceiver transmits a high-frequency transmission signal for a medium to long-distance range that is modulated according to a code (code sequence) having a large pulse code length. In this manner, different kinds of transmission pulses can be used according to the distance range of a measurement subject, whereby pulse echoes due to a nearby, fast-moving target can be reduced.
The testing instruments disclosed in Patent document 2 and 3 are known as testing instruments for completely eliminating interfering echoes by transmission signals generated using a complementary code.
This type of testing instrument generates plural transmission signals having a prescribed code length using complementary code sequences or plural auxiliary sequences and drives plural probes in predetermined order by these transmission signals. The testing instrument generates reference signals according to the code sequences by a reference signal generator. Furthermore, the testing instrument performs correlation processing on plural echoes received by the plural probes by a correlator using the plural reference signals, and adds processing results together. As a result, interfering echoes can be prevented from affecting a test result and the signal-to-noise ratio can be increased.